The present invention relates to a switching power supply circuit provided as a power supply in various electronic apparatus.
Switching power supply circuits employing switching converters such for example as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Accordingly, various switching power supply circuits using various resonance type converters have been previously proposed by the present applicant. A resonance type converter can readily obtain high power conversion efficiency, and achieve low noise because the resonance type converter forms a sinusoidal waveform in switching operation. The resonance type converter has another advantage of being able to be formed by a relatively small number of parts.
FIG. 18 is a circuit diagram showing an example of configuration of a switching power supply circuit that can be formed on the basis of an invention of Japanese Patent Publication No. 2955582 previously proposed by the present applicant. This power supply circuit employs a self-excited current resonance type converter.
The switching power supply circuit shown in the figure is provided with a voltage doubler rectifier circuit formed by rectifier diodes Di1 and Di2 and smoothing capacitors Ci1 and Ci2 as a rectifying and smoothing circuit for receiving an alternating input voltage VAC. The voltage doubler rectifier circuit generates a rectified and smoothed voltage Ei equal to twice the alternating input voltage VAC across the serially connected smoothing capacitors Ci1 and Ci2.
The switching converter of the power supply circuit is connected such that two switching devices Q1 and Q2 are coupled by half-bridge coupling, and inserted between a node on the positive electrode side of the smoothing capacitor Ci1 and a ground, as shown in FIG. 18. In this case, a bipolar transistor (BJT; junction transistor) is employed as the switching devices Q1 and Q2.
An orthogonal type control transformer PRT (Power Regulating Transformer) is provided to drive the switching devices Q1 and Q2 and effect constant-voltage control as later described.
The orthogonal type control transformer PRT is formed as an orthogonal type saturable reactor in which driving windings NB1 and NB2 and a resonance current detecting winding ND for detecting resonance current are wound as shown in FIG. 22, and a control winding NC is wound in a direction orthogonal to these windings.
An isolation converter transformer PIT1 (Power Isolation Transformer) transmits a switching output of the switching devices Q1 and Q2 to a secondary side.
As shown in FIG. 20, the isolation converter transformer PIT1 has an E-E-shaped core formed by combining E-shaped cores CR1 and CR2 of for example a ferrite material in such a manner that magnetic legs of the core CR1 are opposed to magnetic legs of the core CR2, and has a primary winding N1 and a secondary winding N2 (N3) wound around a central magnetic leg of the E-E-shaped core in a state of being divided from each other by a dividing bobbin B. In this case, the primary winding N1 and the secondary windings N2 and N3 are each formed by winding a litz wire of about 60 mmxcfx86 around the dividing bobbin B.
In this case, a gap G of 0.5 mm to 1.0 mm is formed in the central magnetic leg of the E-E-shaped core, whereby a state of loose coupling at a coupling coefficient kxe2x89xa00.85, for example, is obtained between the primary winding N1 and the secondary windings N2 and N3.
One end of the primary winding N1 of the isolation converter transformer PIT1 is connected to a node (switching output point) of an emitter of the switching device Q1 and a collector of the switching device Q2 via the resonance current detecting winding ND, and thereby obtains the switching output. Another end of the primary winding N1 is connected to the primary-side ground via a primary-side series resonant capacitor Cr1 formed by a film capacitor, for example.
A primary-side parallel resonant capacitor Cr2 for primary-side partial voltage resonance is connected in parallel with the collector and emitter of the switching device Q2. The primary-side parallel resonant capacitor Cr2 is provided for ZVS (Zero Voltage Switching) operation and ZCS (Zero Current Switching) operation of the switching devices Q1 and Q2.
The secondary windings N2 and N3 are wound independently of each other on the secondary side of the isolation converter transformer PIT1 in FIG. 18. The secondary winding N2 is connected with a bridge rectifier diode DBR and a smoothing capacitor C01, whereby a direct-current output voltage E01 is generated. The secondary winding N3 is provided with a center tap. The secondary winding N3 is connected with rectifier diodes D01 and D02 and a smoothing capacitor C02 as shown in the figure, whereby a full-wave rectifier circuit formed by the rectifier diodes D01 and D02 and smoothing capacitor C02 generates a direct-current output voltage E02.
In this case, the direct-current output voltage E01 is also inputted from a branch point to a control circuit 1.
The control circuit 1 for example supplies, as a control current, a direct current whose level is changed according to level of the secondary-side direct-current output voltage E01 to the control winding NC of the orthogonal type control transformer PRT, and thereby effects constant-voltage control.
FIG. 19 is a circuit diagram showing an example of configuration of another power supply circuit that can be formed on the basis of an invention previously proposed by the present applicant. The same parts as in the power supply circuit shown in FIG. 18 are identified by the same reference numerals, and their description will be omitted.
The power supply circuit shown in FIG. 19 is also provided with a current resonance type converter in which two switching devices Q11 and Q12 are coupled by half-bridge coupling. However, a driving system for the current resonance type converter is an external excitation system. In this case, a MOS-FET or an IGBT (Insulated Gate Bipolar Transistor) is used as the switching devices Q11 and Q12.
In this case, a rectifying and smoothing circuit formed by a bridge rectifier circuit Di and a smoothing capacitor Ci rectifies and smoothes an alternating input voltage VAC of a commercial alternating-current power supply AC, and thereby generates a direct-current input voltage equal to a peak value of the alternating input voltage VAC multiplied by unity, for example.
Gates of the switching devices Q11 and Q12 are connected to an oscillating and driving circuit 11. The switching device Q11 has a drain connected to a positive electrode of the smoothing capacitor Ci, and a source connected to a primary-side ground via a primary winding N1 and a primary-side series resonant capacitor Cr1. The switching device Q12 has a drain connected to the source of the switching device Q11, and a source connected to the primary-side ground.
Also in this case, a primary-side parallel resonant capacitor Cr2 for primary-side partial voltage resonance is connected in parallel with the drain and the source of the switching device Q12.
Further, a clamp diode DD1 is connected in parallel with the drain and the source of the switching device Q11, and a clamp diode DD2 is connected in parallel with the drain and the source of the switching device Q12.
The switching devices Q11 and Q12 are driven by the oscillating and driving circuit 11 for switching operation described earlier with reference to FIG. 18.
Specifically, a control circuit 2 in this case supplies a current or a voltage varied in level according to variation in a direct-current output voltage E01 to the oscillating and driving circuit 11 on the primary side via a photocoupler PC. In order to stabilize the direct-current output voltage E01, the oscillating and driving circuit 11 outputs a switching driving signal (voltage) varied in cycle according to the level of the output from the control circuit 2 to the gates of the switching devices Q11 and Q12 alternately. Thereby switching frequency fs of the switching devices Q11 and Q12 is varied.
In this case, the oscillating and driving circuit 11 is supplied with a starting voltage via a starting resistance RS, and supplied with a smoothed output, as a driving voltage therefor, obtained by smoothing an output of a winding N4 wound additionally on the primary side of an isolation converter transformer PIT2 by a capacitor C1.
As an example of characteristics of the power supply circuit shown in FIG. 18, FIG. 21 shows characteristics of variations in AC-to-DC power conversion efficiency xcex7ACxe2x86x92DC, switching frequency fs, and the period TON of the switching device Q2 when load power Po of the secondary-side direct-current output voltage E01 is varied from 0 W to 200 W.
FIG. 21 shows characteristics when the number of turns of each of the primary winding Ni and the secondary winding N2 of the isolation converter transformer PIT1 is selected to be 45 T (turns), 0.056 xcexcF is selected for the primary-side series resonant capacitor Cr1, and 330 pF is selected for the primary-side parallel resonant capacitor Cr2 so as to correspond to conditions of an AC 200 V system.
As shown in FIG. 21, the switching frequency fs of the power supply circuit shown in FIG. 18 is controlled to be lowered as the load power Po is increased. Also, as the load power Po is increased, the period TON during which the switching device Q2 is on is controlled to be lengthened.
The power conversion efficiency xcex7ACxe2x86x92DC in this case is about 91.8% at a load power Po of 200 W, and is about 92.4% at a load power Po of 150 W. Thus, the best efficiency is obtained at the load power Po of 150 W.
The AC-to-DC power conversion efficiency xcex7ACxe2x86x92DC of the power supply circuit shown in FIG. 18 when the voltage doubler rectifier circuit as shown in FIG. 18 rectifies the commercial alternating voltage is about 92%, and the AC-to-DC power conversion efficiency xcex7ACxe2x86x92DC of the power supply circuit shown in FIG. 19 when the full-wave rectifier circuit as shown in FIG. 19 rectifies the commercial alternating voltage is about 90%. It is desirable that power loss resulting from such AC-to-DC power conversion be as small as possible.
As a means for improving the AC-to-DC power conversion efficiency xcex7ACxe2x86x92DC, it is conceivable that the gap G is not formed in the central magnetic leg of the E-E-shaped core forming the isolation converter transformer PIT1 or PIT2.
However, in the case of the power supply circuit shown in FIG. 18, for example, a range of the switching frequency fs of the switching devices Q1 and Q2 needs to be set so that a predetermined secondary-side direct-current output voltage E01 (for example 135 V) is obtained even when the alternating input voltage VAC is a minimum alternating input voltage (for example 90 V) and the load power Po is a maximum load power (for example 200 W). That is, a range of series resonance frequency fo determined by leakage inductance of the primary winding N1 of the isolation converter transformer PIT1 and capacitance of the primary-side series resonant capacitor Cr1 needs to be determined in consideration of the case where the alternating input voltage VAC is the minimum alternating input voltage.
Thus, in the power supply circuit shown in FIG. 18, a small capacitance value must be selected as the capacitance value of the primary-side series resonant capacitor Cr1, and unless a gap G of about 0.5 mm to 1.0 mm is formed in the central magnetic leg of the isolation converter transformer PIT1, the switching devices Q1 and Q2 cannot be operated stably by ZVS and ZCS.
In addition, the gap G in the central magnetic leg of the isolation converter transformer PIT1 or PIT2 as shown in FIG. 20 is formed by grinding a ferrite core. Therefore a grinding process is required, and increases cost correspondingly.
Further, the primary winding N1 and the secondary winding N2 in the proximity of the gap of the isolation converter transformer PIT1 or PIT2 increase the temperature due to an eddy current loss caused by a fringe magnetic flux. Further, since the isolation converter transformers PIT1 and PIT2 are a loosely coupled transformer, measures for shielding against leakage flux such as providing a short ring formed by a copper plate around the periphery of the transformer are required.
Accordingly, in view of the above problems, a switching power supply circuit according to the present invention is comprised as follows.
The switching power supply circuit according to the present invention includes: switching means formed by half-bridge coupling of two switching devices, for performing switching operation on a direct-current input voltage; an isolation converter transformer having a primary winding and a secondary winding formed around a magnetic core having no gap formed therein, for transmitting an output of the switching means obtained in the primary winding to the secondary winding; and a primary-side series resonant circuit formed at least by a leakage inductance component including the primary winding of the isolation converter transformer and capacitance of a primary-side series resonant capacitor connected in series with the primary winding, for converting the switching operation of the switching devices into current resonance type operation. The switching power supply circuit further includes: a primary-side partial resonance capacitor connected in parallel with one of the two switching devices, for effecting partial resonance in timing corresponding to a turn-off time of the two switching devices; switching driving means for applying a switching driving signal to the two switching devices for switching operation; a series circuit connected in parallel with the primary-side series resonant capacitor, and formed at least by a series connection of an auxiliary switching device for performing switching operation so as to have an on period in correspondence with timing of turning off of the switching device with which the primary-side partial resonance capacitor is connected and a capacitor for which more than a predetermined capacitance value is selected; and direct-current output voltage generating means for receiving an alternating voltage obtained at the secondary winding of the isolation converter transformer as an input and generating a predetermined secondary-side direct-current output voltage.
With the above configuration, since no gap is formed in the isolation converter transformer, AC-to-DC power conversion efficiency is improved and leakage magnetic flux is reduced. In this case, the auxiliary switching device of the series circuit connected in parallel with the primary-side series resonant capacitor is operated only during a period when the switching device connected in parallel with the primary-side partial resonance capacitor is on. During this period, the capacitor of the series circuit is connected in parallel with the primary-side series resonant capacitor. Thus, even when the gap of the isolation converter transformer is reduced to zero, the switching devices can be operated stably by zero voltage switching and zero current switching.